ADP3031 pra

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PRELIMINARY TECHNICAL DATA
a
2 MHz PWM Boost
Switching Regulator
Preliminary Technical Data
ADP3031
FEATURES
Up to 2 MHz PWM Frequency
Fully Integrated 1.5 A Power Switch
3% Output Regulation Accuracy
Adjustable Output Voltage from 3 V to 12 V
Simple Compensation
Small Inductor and MLC Capacitors
90% Efficiency
Under-voltage Lockout
Shutdown
Functional Block Diagram
COMP
IN
1
6
ERROR
AMP
ADP3031
REF
g
m
BIAS
FB
2
F/F
5
SW
RAMP
GEN
R
Q
S
APPLICATIONS
TFT LCD Bias Supplies
Portable Electronic Instruments
Industrial Systems
COMPARATOR
DRIVER
RT
7
OSC
SD
3
CURRENT
SENSE
AMPLIFIER
GENERAL DESCRIPTION
The ADP3031 is a high frequency, step-up DC-DC
switching regulator capable of 12 V boosted output voltage
in a space saving MSOP-8 package. It provides high
efficiency, low-noise operation, and is easy to use. Ca-
pable of operating up to 2 MHz, the high switching fre-
quency and PWM current mode architecture allow for
excellent transient response, ease of noise filtering, and also
small, cost-saving external inductive and capacitive compo-
nents. The current limit and the power switch are inte-
grated completely on chip.
Capable of operating from 2.5 V to 5.5 V input, the
ADP3031 is ideal for Thin-Film Transistor (TFT) Liquid
Crystal Display (LCD) module applications, where local
point-of-use power regulation is required. Featuring an
adjustable output that supports voltages down to 3 V, the
ADP3031 is ideal to generate today’s low voltage rails,
providing the optimal solution in its class for delivering
power efficiently, responsively, and simply with minimal
printed circuit board area.
The device is specified over the industrial temperature
range of -40°C to +85°C.
8
GND
4
PGND
REV. PrA 2/8/02
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties which may result from
its use. No license is granted by implication or otherwise under any patent or
patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel:781/329-4700
www.analog.com
Fax:781/326-8703
©ANALOG DEVICES, INC., 2002
APPLICATIONS
ADP3031–SPECIFICATIONS
1
Parameter
PRELIMINARY TECHNICAL DATA
(VIN = +3.3 V, T
A
= –40°C to 85°C, unless otherwise noted)
Symbol
Conditions
Min
Typ
Max
Units
SUPPLY
Input Voltage
V
IN
2.5
5.5
V
Quiescent Current
Switching State
3
I
QSW
f = 600 kHz, light load
2
5
mA
Non-switching State
I
Q
300
500
µA
Shutdown
I
SD
10
µA
ERROR AMPLIFIER
Feedback Voltage
V
FB
TBD 1.233 TBD V
Line Regulation
V
IN
= 2.5 V to 5.5 V
–0.1
+0.1
%/V
Bias Current
100
nA
Overall Regulation
Line, Load, Temp
–3
3
%
OUTPUT SWITCH
On Resistance
R
ON
at 1.5 A, V
IN
= 3.3 V
200
mΩ
Output Load Current
2
I
LOAD
continuous operation
200
300
mA
V
IN
= 3.3 V,V
OUT
=10 V
Leakage Current
V
SWITCH
= 12 V
5
µA
Efficiency
2
I
LOAD
= 200 mA,
85
%
V
OUT
= 10 V, f = 600 kHz
Efficiency
2
I
LOAD
= 100 mA,
90
%
V
OUT
= 10 V, f = 600 kHz
OSCILLATOR
Oscillator Frequency
FOSC
RT = Open
0.5
0.6
0.7
MHz
RT = GND
1.7
2
2.3
MHz
Maximum Duty Cycle
D
MAX
COMP = open, FB = 1 V
80
85
%
SHUTDOWN
Shutdown Input Voltage Low
0.4
V
Shutdown Input Voltage High
1.0
V
CURRENT LIMIT
Peak Switch Current
I
CLSET
1.5
1.8
A
COMPENSATION
Transconductance
g
m
100
µA/V
Gain
A
V
1000
V/V
UNDER-VOLTAGE LOCKOUT
UVLO Threshold
2.2
2.45
2.5
V
UVLO Hysteresis
100
mV
OUTPUT
Voltage Range
V
OUT
7
12
V
NOTES
1
All limits at temperature extremes are guaranteed via correlation and characterization using standard Statistical Quality Control (SQC).
2
See Figure xx.
This is the average current while switching.
Specifications subject to change without notice.
–2–
REV. PrA
3
PRELIMINARY TECHNICAL DATA
ADP3031
ABSOLUTE MAXIMUM RATINGS*
Input Voltage .............................................. –0.3 V to +6 V
SW Voltage ................................................................. 14 V
COMP Voltage ........................................ –0.3 V to +2.5 V
FB Voltage ............................................... –0.3 V to +1.3 V
SD
Voltage ................................................. –0.3 V to +6 V
RT Voltage .............................................. –0.3 V to +1.2 V
PGND TO GND ................................................ ±200 mV
Operating Ambient Temperature Range ..... –40°C to 85°C
Operating Junction Temperature Range ..... –40°C to 125°C
Storage Temperature Range ................... –65°C to +150°C
θ
JA
Two Layer ..................................................... 206°C/W
θ
JA
Four Layer ..................................................... 142°C/W
Lead Temperature Range (Soldering, 60 sec.) ......... 300°C
*This is a stress rating only; operation beyond these limits can cause the device
to be permanently damaged. Unless otherwise specified, all voltages are
referenced to GND
PIN FUNCTION DESCRIPTIONS
Pin
Mnemonic Function
1
COMP
Compensation Input.
2
FB
Feedback voltage sense input.
3
SD
Shutdown Input.
4
PGND
Power Ground. Ground return for
power transistor.
5
SW
Switching Output.
6
IN
Main Power Supply Input.
7
RT
Frequency Setting Input. A resistor
between this pin and GND sets the
switching frequency of the device.
8
GND
Analog Ground. The control
circuitry is referenced to this ground.
ORDERING GUIDE
Voltage Package Branding
PIN CONFIGURATION
Model
Output Option Information
ADP3031ARM
ADJ
MSOP-8 P8A
COMP
FB
1
2
3
4
8
7
6
5
GND
MSOP8
RT
TOP VIEW
(Not to Scale)
SD
PGND
IN
SW
R
C
IN
C
C
V
OUT
COMP
IN
C
IN
1
6
ERROR
AMP
ADP3031
L1
R
1
REF
FB
g
m
BIAS
2
COMPARATOR
SW
D1
R
2
5
V
OUT
F/F
R
Q
C
OUT
RAMP
GEN
S
DRIVER
RT
7
OSC
R
T
SD
3
CURRENT
SENSE
AMPLIFIER
GND
8
4
PGND
Figure 1.Typical Application
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although the device features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
REV. PrA
–3–
PRELIMINARY TECHNICAL DATA
ADP3031
THEORY OF OPERATION
The ADP3031 is a boost converter driver which stores
energy from an input voltage in an inductor, and delivers
that energy, augmented by the input, to a load at a higher
output voltage. It includes a voltage reference and error
amplifier to compare some fraction of the load voltage to
the reference, and amplify any difference between them.
The amplified error signal is compared to a dynamic signal
produced by an internal ramp generator incorporating
switch current feedback. The comparator output timing
sets the duty ratio of a switch driving the inductor to
maintain the desired output voltage.
Referring to Figure 1, a typical application will power both
the IC and the inductor from the same input voltage. The
on chip MOSFET will be driven on, pulling pin SW close
to PGND. The resulting voltage across the inductor will
cause its current to increase aproximately linearly, with
respect to time.
When the MOSFET switch is turned off the inductor
current cannot drop to zero, and so this current drives the
SW node capacitance rapidly positive until the diode
becomes forward biased. The inductor current will now
begin to charge the load capacitor, causing a slight increase
in output voltage. Generally, the load capacitor is made
large enough that this increase is very small during the time
the switch is off. During this time inductor current is also
delivered to the load. In steady state operation, the induc-
tor current will exceed the load current, and the excess will
be what charges the load capacitor. The inductor current
will fall during this time, though not necessarily to zero.
During the next cycle, initiated by the on-chip oscillator,
the switch will again be turned on so that the inductor
current will be ramped back up. The charge on the load
capacitor will provide load current, during that interval.
The remainder of the chip is arranged to control the duty
ratio of the switch, to maintain a chosen output voltage
despite changes in input voltage or load current.
The output voltage is scaled down by a resistor voltage
divider and presented to the g
m
amplifier. This amplifier
operates on the difference between an on-chip reference
and the voltage at the FB pin so as to bring them to bal-
ance. This will be when the output voltage equals the
reference voltage, multiplied by the resistor voltage divider
ratio.
The g
m
amplifier drives an internal comparator, which has
at its other input a positive going ramp produced by the
Oscillator and modified by the current sense amplifier. The
MOSFET switch is turned on as the modified ramp voltage
rises. When this voltage exceeds the output of the g
m
amplifier the comparator will turn off the switch, by
resetting the flip-flop, previously set by the oscillator. The
output of the flip-flop is buffered by a high current driver
which turned on the MOSFET switch at the beginning of
the Oscillator cycle.
In the steady state with constant load and input voltage, the
current in the inductor will cycle around some average
current level. The increasing ramp of current will depend
on input voltage and t1, the switch on-time, while the
decreasing ramp will depend on the difference between
input and output voltage and t2, the remainder of the cycle.
In order for the peaks of these two ramps to be equal and
opposite to maintain steady state we can say that t1*V
IN
will
equal t2*(V
OUT
-V
IN
), if we neglect the effect of resistance in
the inductor and switch, and the forward voltage drop of
the diode. From this equality we can derive t1/T=1-V
IN
/
V
OUT
, where T is the period of a cycle, t1+t2. This result
gives us the switch duty ratio, t1/T in terms of the input and
output voltages.
In practice the duty ratio will need to be slightly higher than
this calculation. Because of series resistance in the inductor
and the switch, the voltage across the actual inductance is
somewhat less than applied V
IN
, and the actual output
voltage is less than our aproximation by the amount of the
diode forward voltage drop. However, the feedback
control within the ADP3031 will adjust the duty ratio to
maintain the output voltage. Changes in load current and
input voltage are also accomodated by the feedback
control.
Changes in load current alone require a change in duty
ratio, in order to change the average inductor current. But
once the inductor current adapts to the new load current,
the duty ratio should return to nearly its original value, as
we see from the duty cycle calculation which depends on
input and output voltages, but not on current. Increasing
the switch duty ratio initially reduces the output voltage,
until the average inductor current increases enough to
offset the reduction of the t2 interval. By limiting the duty
ratio we prevent this effect from regeneratively increasing
the duty ratio to 100%, which would cause the output to
fall and the switch current to rise without limit. The duty
ratio is limited to about 80% by the design of the Oscillator
and an additional flip-flop reset.
A comparator compares the current sense amplifier output
to a factory set limit which resets the flip-flop, turning off
the switch. This prevents runaway or overload conditions
from damaging the switch and reflecting fault overloads
back to the input. Of course, the load is directly connected
to the input by way of the diode and inductor, so protection
against short circuited loads must be done at the power
input.
The g
m
amplifier has high voltage gain, to insure the output
voltage accuracy and invariance with load and input
voltage. However, because it is a g
m
amplifier with a
specified current response to input signal voltages, its high
frequency response can be controlled by the compensation
impedance. This permits the high frequency gain of the g
m
amplifier to be optimized for the best compromise between
speed of response and frequency stability.
The stable closed loop bandwidth of the system can be
extended by the current feedback shown. A signal repre-
senting the magnitude of the switch current is added to the
ramp. This dynamically reduces the duty ratio, as the
current in the inductor increases, until the g
m
amplifier
restores it, improving the closed loop frequency stability.
–4–
REV. PrA
PRELIMINARY TECHNICAL DATA
ADP3031
The ADP3031 is intended to operate over a range of
frequencies, set by the RT pin. If the pin is open, the
oscillator runs at its lowest frequency: if the pin is
“grounded” it runs at its highest frequency. A resistor from
RT to ground can be used to set intermediate operating
frequencies.
Because of the large currents which flow in the main
MOSFET switch, it is provided with a separate PGND
return to the negative supply terminal, to avoid corrupting
the small signal return, GND, that can be used as a sense
line at the output load point.
APPLICATION INFORMATION
Frequency Selection
The ADP3031's frequency can be user selected to operate at either 600
KHz or 2 MHz and programmable by setting the RT pin. Tie RT to
GND for 2 MHz operation. For 600 KHz operation, float the RT pin.
The nominal resistance at the RT pin to get a switching
frequency, f
SW
, is given by:
Inductor Selection
For most of the applications, the inductor used with the
ADP3031 should be in the range of 2 µH to 22 µH. Several
inductor manufacturers are listed in Table 1. When
selecting an inductor, it is important to make sure that the
inductor used with the ADP3031 is able to handle the peak
current without saturation and that the peak current is
below the current limit of the ADP3031.
As a rule, powdered iron cores saturate softly, whereas
Ferrite cores saturate abruptly. Open drum core inductors
tend to saturate gradually, are low cost and are small in
size, making these types of inductors attractive in many
applications. However, care must be exercised in their
placement because they have high magnetic fields. In
applications that are sensitive to magnetic fields, shielded
geometries are recommended.
In addition, inductor losses must be considered. Both core
and copper losses contribute to loss in converter efficiency.
To minimize core losses, look for inductors rated for
operation at high switching frequencies. To minimize
copper losses, it is best to use low dc resistance inductors.
Typically, it is best to use an inductor with a dc resistance
lower than 20 mΩ per µH.
The inductor value can be estimated using the following:
L = (V
OUT
- V
IN
) × M
SLOPE
Where M
SLOPE
= scaling factor for proper slope compen-
sation.
RT (Ω ) = 320,000 x (2,000,000 - f
SW
)/(3.6667 x f
SW
–
2,000,000)
(1)
Output Voltage
The ADP3031 features an adjustable output voltage range of
V
IN
to 12 V. The output voltage is fed back to the ADP3031 via
resistor dividers R1 and R2 (Figure 1.). The feedback voltage
is 1.233 V, so the output voltage is set by the formula:
V
OUT
= 1.233 ×
( 1+ R1/R2)
(2)
1.456
M
Since the feedback bias current is 100 nA maximum, R2 may
have a value up to 100 KΩ
SLOPE
f
SW
with minimum error due to the bias
current.
Choose the closest standard inductor value as a starting
point.
TABLE 1. INDUCTOR MANUFACTURERS
Max DC
Max DCR
Height
Part
L(µH)
Current
m
Ω Ω
(mm)
Vendor
CMD4D11-2R2MC
2.2
0.95
116
1.2
Sumida
CMD4D11-4R7MC
4.7
0.75
216
1.2
847-956-0666
CDRH4D28-100
10
1.00
128
3.0
www.sumida.com
CDRH5D18-220
22
0.80
290
2.0
CR43-4R7
4.7
1.15
109
3.5
CR43-100
10
1.04
182
3.5
DS1608-472
4.7
1.40
60
2.9
Coilcraft
DS1608-103
10
1.00
75
2.9
847-639-6400 www.coilcraft.com
D52LC-4R7M
4.7
1.14
87
2.0
Toko
D52LC-100M
10
0.76
150
2.0
847-297-0070 www.tokoam.com
REV. PrA
–5–
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